Method and apparatus to reduce audio frequencies in a switching power supply

ABSTRACT

An example controller for use in a power supply regulator includes a switch signal generator, a modulation circuit, and a multi-cycle modulator circuit. The modulation circuit modulates the period of a modulation switching signal when an equivalent switching frequency is greater than a reference frequency and fixes the switching period when the equivalent switching frequency is less than the reference frequency. The multi-cycle modulator circuit enables the switch signal generator to provide a switch signal uninterrupted if the equivalent switching frequency is greater than the reference frequency and disables the switch signal generator for a first time period and then enables the switch signal generator for a second time period when the equivalent frequency is less than the reference frequency. The multi-cycle modulator circuit varies the first time period to regulate the output.

REFERENCE TO PRIOR APPLICATIONS

This application is a continuation of U.S. application Ser. No.13/102,986, filed May 6, 2011, now pending, which is a continuation ofU.S. application Ser. No. 12/767,564, filed Apr. 26, 2010, now U.S. Pat.No. 7,965,524, which is a continuation of U.S. application Ser. No.12/276,222, filed Nov. 21, 2008, now U.S. Pat. No. 7,733,673, which is acontinuation of U.S. application Ser. No. 11/543,518, filed Oct. 4,2006, now U.S. Pat. No. 7,471,530. U.S. application Ser. No. 13/102,986,and U.S. Pat. Nos. 7,965,524, 7,733,673 and 7,471,530 are herebyincorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to power supplies and, morespecifically, the present invention relates to a switching regulator.

2. Background Information

Electronic devices use power to operate. Switched mode power suppliesare commonly used due to their high efficiency and good outputregulation to power many of today's electronic devices. In a knownswitching power supply, a low frequency (e.g. 50 or 60 Hz mainsfrequency), high voltage alternating current (AC) is converted to highfrequency (e.g. 30 to 300 kHz) AC, using a switched mode power supplycontrol circuit. This high frequency, high voltage AC is applied to atransformer to transform the voltage, usually to a lower voltage, and toprovide safety isolation. The output of the transformer is rectified toprovide a regulated DC output, which may be used to power an electronicdevice. The switched mode power supply control circuit usually providesoutput regulation by sensing the output and controlling it in a closedloop.

The design of the switching power supply is a compromise amongconflicting requirements of efficiency, size, weight, and cost. Theoptimal solution that delivers the rated output power usually sets theswitching frequency much higher than 20 kHz, outside the range of humanhearing.

Regulatory requirements call for power supplies to operate at highefficiency at low loads such as standby loads and consume very low powerat no load. When a power supply delivers much less than its rated power,the energy lost within the power supply is dominated by losses from theaction of switching. Therefore, it is necessary for the power supply tooperate at lower switching frequencies when the output power is low toreduce the dominant losses. The switching frequency may be reducedlinearly as the load reduces to maintain high efficiency. The optimalswitching frequency at low power often falls within the band of audiofrequencies below 20 kHz. Switching within the band of audio frequenciescan produce undesirable audio noise in power supply components such astransformers and ceramic capacitors due to mechanical resonances.

A well-known technique to reduce switching losses and improve efficiencyat light loads is to operate the power supply in a burst mode at lightloads. Burst mode operation allows the power supply to switch anuncontrolled number of consecutive switching cycles at a high switchingfrequency followed by a duration of no switching adjusted in a closedloop to regulate the output. Thus, the average switching frequency isreduced to keep the efficiency high at light loads. An undesirableproperty of burst mode switching is that neither the number ofconsecutive high frequency switching cycles in the burst nor the numberof consecutive cycles of no switching between bursts is determined for agiven set of operating conditions. The indeterminate nature of burstmode operation creates the hazard of uncontrolled audio noise. In fact,if the repetition rate of the duration of consecutive switching cyclesfollowed by the duration of no switching is at audio frequency, theaudio noise could be worse than just reducing the frequency linearly asdescribed earlier due to higher audio energy content. Audio noise is amajor drawback of using burst mode operation for improving light loadefficiency and reducing no load consumption.

One of the most troublesome sources of noise in a switching power supplyis the transformer. Commonly used ferrite core transformers in switchingpower supplies tend to have mechanical resonant frequencies in the 8 kHzto 15 kHz range. Some ceramic capacitors, especially the ones used inthe clamp circuit connected to the primary winding of the transformer ina flyback power supply, can also resonate at such audio frequencies.

BRIEF DESCRIPTION OF THE DRAWINGS

Non-limiting and non-exhaustive embodiments of the present invention aredescribed with reference to the following figures, wherein likereference numerals refer to like parts throughout the various viewsunless otherwise specified.

FIG. 1A is a block diagram illustrating generally an example switchingpower supply providing power to a load.

FIG. 1B is a diagram illustrating generally an example of a switchingcurrent waveform of the example switching power supply of FIG. 1A.

FIG. 2 is an example flow diagram illustrating generally a sample flowof operations of an example switching power supply providing power to aload in accordance with the teachings of the present invention

FIG. 3A is an example schematic illustrating generally an examplecontroller of an example switching power supply providing power to aload in accordance with the teachings of the present invention.

FIG. 3B is an example schematic illustrating generally a fixed frequencypeak current PWM modulator and a selector comparator circuit thatselects modes of operation in accordance with the teachings of thepresent invention.

FIG. 4 is another example flow diagram illustrating generally a sampleflow of operations of an example switching power supply providing powerto a load in accordance with the teachings of the present invention

FIG. 5 is another example schematic illustrating generally an examplecontroller of an example switching power supply providing power to aload in accordance with the teachings of the present invention.

FIG. 6A is a diagram illustrating generally an example of magnetic fluxdensity in an energy transfer element of a switching power supplyoperating in a period modulation mode in accordance with the teachingsof the present invention.

FIG. 6B is a diagram illustrating generally an example of partialfrequency spectra in an energy transfer element of a switching powersupply operating at high frequency in accordance with the teachings ofthe present invention.

FIG. 6C is a diagram illustrating generally an example of partialfrequency spectra in an energy transfer element of a switching powersupply operating at low frequency without multi-cycle modulation inaccordance with the teachings of the present invention.

FIG. 7A is a diagram illustrating generally an example of magnetic fluxdensity in an energy transfer element of a switching power supplyoperating in a multi-cycle modulation mode in accordance with theteachings of the present invention.

FIG. 7B is a diagram illustrating generally an example of partialfrequency spectra in an energy transfer element of a switching powersupply operating in a multi-cycle modulation mode in accordance with theteachings of the present invention.

FIG. 8A is a diagram illustrating generally an example of an unmodulatedmagnetic flux density signal in an energy transfer element of aswitching power supply in accordance with the teachings of the presentinvention.

FIG. 8B is a diagram illustrating generally an example of a multi-cyclemodulator signal of a switching power supply in accordance with theteachings of the present invention.

FIG. 8C is a diagram illustrating generally an example relativemagnitudes of the coefficients for positive and negative frequencies ina complex Fourier series representation of a multi-cycle modulatorsignal in accordance with the teachings of the present invention.

FIG. 9 is a diagram illustrating generally how an example unmodulatedswitching signal and an example spectrum of a multi-cycle modulatorsignal are combined to produce an example spectrum of a multi-cyclemodulated signal in accordance with the teachings of the presentinvention.

DETAILED DESCRIPTION

Methods and apparatuses for maintaining high efficiency while reducingaudio frequencies in a switching power supply are disclosed. In thefollowing description numerous specific details are set forth in orderto provide a thorough understanding of the present invention. It will beapparent, however, to one having ordinary skill in the art that thespecific detail need not be employed to practice the present invention.In other instances, well-known materials or methods have not beendescribed in detail in order to avoid obscuring the present invention.

Reference throughout this specification to “one embodiment,” “anembodiment,” “one example,” “an example” or the like means that aparticular feature, structure or characteristic described in connectionwith the embodiment or example is included in at least one embodiment orexample of the present invention. Thus, appearances of the phrases “inone embodiment,” “in an embodiment,” “one example” or “an example” invarious places throughout this specification are not necessarily allreferring to the same embodiment or example. Furthermore, the particularfeatures, structures or characteristics may be combined in any suitablecombinations and/or subcombinations in one or more embodiments orexamples. In addition, it is appreciated that the figures providedherewith are for explanation purposes to persons ordinarily skilled inthe art and that the drawings are not necessarily drawn to scale.

As will be discussed, examples of switching a switch of a power supplyare disclosed that greatly reduce the mechanical excitation of powersupply components at audio frequencies while maintaining the benefits ofswitching at an equivalent audio frequency. According to variousexamples, switching cycles with short periods are gathered into groupsthat are separated by intervals of no switching. Regulation of an outputof the switching power supply is accomplished by controlling theduration of a group and the duration of no switching between groups ofswitching cycles of the switch in accordance with the teachings of thepresent invention.

To illustrate, FIG. 1A shows generally a functional block diagram of anexample switching power supply regulator 100. It is noted that theexample switching power supply regulator 100 shown in FIG. 1A is aflyback regulator, which is one example of a switching power supplytopology that may benefit from the teachings of the present invention.However, it is appreciated that other known topologies andconfigurations of switching power supply regulators may also benefitfrom the teachings of the present invention.

The example switching power supply regulator 100 in FIG. 1A providesoutput power to a load 165 from an unregulated input voltage VIN 105.The input voltage VIN 105 is coupled to an energy transfer element T1125 and a switch S1 120. In the example of FIG. 1A, the energy transferelement T1 125 is a transformer with two windings. A clamp circuit 110is coupled to the primary winding of the energy transfer element T1 125to control the maximum voltage on the switch S1 120. Switch S1 120 isopened and closed in response to switch signal 112 received fromcontroller circuit or controller 145. In the illustrated example, switchS1 120 is a transistor and controller 145 includes integrated circuitsand/or discrete electrical components. In operation, the switching ofswitch S1 120 produces a pulsating current in the rectifier D1 130 thatis filtered by capacitor C1 135 to produce a substantially constantoutput voltage Vo or output current I_(O) at the load 165.

The output quantity to be regulated is U_(O) 150, that in general couldbe an output voltage V_(O), an output current I_(O), or a combination ofthe two. A feedback circuit 160 is coupled to sense the output quantityU_(O) 150 from the output of the power supply to produce a feedbacksignal U_(FB) 155 that is an input to the controller 145. Another inputto the controller 145 is the current sense signal 140 that senses acurrent I_(D) 115 in switch S1 120. Any of the many known ways tomeasure a switched current, such as for example a current transformer,or for example the voltage across a discrete resistor, or for examplethe voltage across a transistor when the transistor is conducting, maybe used to measure current I_(D) 115.

In the example of FIG. 1A, the controller 145 operates switch S1 120 inresponse to system inputs to substantially regulate output quantityU_(O) 150 to its desired value. The output of the power supply istherefore regulated in a closed loop. Controller 145 typically includesan oscillator that defines a switching cycle with a period T_(S).Regulation may be accomplished by control of one or more switchingparameters that determine the amount of energy transferred from theinput to the output of the power supply regulator 100.

FIG. 1B is a graph showing generally current I_(D) 115 as a function oftime. In one example, switch S1 120 turns on at the beginning of eachswitching cycle. Switch S1 120 conducts until current I_(D) 115 reachesa threshold I_(MAX). In one example, control of the maximum currentI_(MAX) at a constant switching cycle period T_(S) (a constant switchingfrequency) maintains an average of the current I_(D) 115 at a valuerequired to regulate the output quantity U_(O) 150. The regulation isthus accomplished by a fixed frequency pulse width modulation techniqueknown as fixed frequency peak current pulse width modulation (PWM) or asfixed frequency current mode control. In another example, regulation maybe accomplished by changing the on time of the switch at fixedfrequency. This mode of control is known as fixed frequency voltage modecontrol or fixed frequency voltage mode PWM. Although the association tovoltage mode originated from its use to regulate an output voltage of apower supply, the term continues to be used to describe the same mode ofcontrol even when the regulated output is a current, or a combination ofa voltage and a current. The term fixed frequency PWM covers bothcurrent mode and voltage mode controls operating at fixed frequency. Inanother example, regulation may be accomplished with another form of PWMknown as period modulation or switching frequency modulation, where thecurrent in the switch reaches a fixed peak value in every switchingcycle and the period Ts is adjusted. At light loads that correspond tolow average values of the current I_(D), it is desired to operate atlower switching frequencies, usually within the range of audiofrequencies, to reduce switching losses in the power supply. All threeof the above modulation techniques involve modulation of the duty cycleand, therefore, are referred to by the general term “duty cyclemodulation”.

In practical applications, the average value of current I_(D) 115required to regulate the output quantity U_(O) 150 may be an averageover not one single switching cycle, but over a time that includes manyswitching cycles. The many switching cycles need not have the sameperiod value. Many different sequences of switching cycles that containdifferent period values can produce an average current I_(D) 115 thatregulates the output quantity U_(O) 150 within its specified limits.

It is desirable to control the period values and the sequences ofswitching cycles to avoid frequencies that can produce audio noise. Aswill be demonstrated later in this disclosure, the magnitudes offrequencies that can produce audio noise may be substantially reduced bygathering the switching cycles into groups of multiple cycles that aremodulated to regulate an output in accordance with the teaching of thepresent invention.

FIG. 2 is an example flow diagram that describes generally a method tocontrol a power supply according to the teachings of the presentinvention. Starting in block 205, sensing of power supply output(s)occurs at block 210.

Next, in block 215, the controller uses information sensed from thepower supply output(s) to estimate the peak current I_(PK) required toregulate the output with a fixed switching cycle period T_(S). Next, theestimate of peak current I_(PK) is compared to a minimum peak currentI_(PKMIN) at a decision block 220. In one example, the minimum peakcurrent I_(PKMIN) corresponds to a light load that requires a reductionin average switching frequency to reduce switching losses.

The estimate of peak current I_(PK) and comparison to the minimum peakcurrent I_(PKMIN) do not need to be precise or the result of numericalcomputation. In one example, the peak current I_(PK) is estimated fromthe magnitude of a feedback signal. The premise of the estimate in oneexample is that a large feedback signal implies a light load thatdemands a switching frequency within the range of audio frequencies,which can cause objectionable levels of audible noise, to meet therequirements of high efficiency. Under those conditions, decision block220 finds that I_(PKMIN)>I_(PK), and the flow is directed to block 230.Peak current I_(PK) is fixed at the value I_(PKMIN) and the period ofthe switching cycle T_(S) is fixed at the value T_(SG) in block 230.Regulation is then accomplished with multi-cycle modulation in a block235.

As shown in block 235, multi-cycle modulation accomplishes regulation ofthe output by gathering N periods of switching in the switch signal,each having period T_(SG), followed by no switching for M periods ofT_(SG). In the example, N and M are modulated, restricted respectivelyto minimum values N_(MIN) and M_(MIN). The resulting pattern ofswitching and no switching in the switch signal gives the same averagecurrent I_(D) 115 in FIG. 1A which has an uninterrupted switching at afixed switching cycle period that corresponds to a higher switchingfrequency that would produce greater switching loss and reducedefficiency. In the example, the pattern of switching with multi-cyclemodulation in accordance with the teachings of the present inventiongenerates substantially less audio noise than an ungrouped pattern withordinary peak current pulse width modulation.

If the decision block 220 finds that I_(PKMIN)≦I_(PK), then the flow isdirected to a block 225 so that regulation is accomplished with peakcurrent PWM at fixed switching frequency instead of multi-cyclemodulation. In one embodiment, I_(PKMIN) is set at approximately 25% ofthe peak current at maximum output power of the power supply. In oneembodiment, the fixed switching frequency is 30 kHz. In one example,multi-cycle modulation may be used to regulate the delivery of lowoutput power and fixed frequency peak current PWM also known as fixedfrequency peak current mode control may be used to regulate the deliveryof higher output power in accordance with the teachings of the presentinvention.

Thus, in one example, dependent upon the relationship between peakswitch current I_(PK) and minimum peak switch current I_(PKMIN), theswitch may be controlled to conduct according to a first or a secondoperating mode. In the first operating mode, a switch is controlled toconduct within a fixed period of the switching cycle and the peak switchcurrent within the fixed period of the switching cycle is adjusted toregulate the output of a power supply. Alternatively, in a secondoperating mode, the switch may be controlled to conduct within aswitching cycle having a fixed period, one group of consecutiveswitching cycles separated from a next group of consecutive switchingcycles by a time of no switching, the time of no switching is adjustedto regulate the output of the power supply. In the example, the time ofno switching may be adjusted so that the frequency of the switchingcycle falls outside an audible frequency range. In the example, thegroup of switching cycles includes at least two consecutive switchingcycles.

In one example, the fixed frequency voltage mode PWM could be usedinstead of peak current mode PWM to regulate delivery of higher power inaccordance with the teachings of the present invention. In this example,the decision to use PWM versus multi-cycle modulation would be based onthe on time of the switch rather than peak current that is needed tokeep the output in regulation. For example, when the on time decreasesin response to the feedback signal and crosses a threshold that wouldindicate a light load, the operation is changed from fixed frequencyvoltage mode PWM to multi-cycle modulation.

FIG. 3A illustrates generally an example schematic 345 of a multi-cyclemodulator 340 included in a controller for a power supply that uses themethod illustrated in FIG. 2. In the example, a feedback control block302 is coupled to receive a sense signal 350 that is related to theoutput(s) of the power supply. Sense signal 350 may represent acombination of one or more parameters such as output voltage, outputcurrent, current in the switch, timing of the switch, power supply inputvoltage, enable signals, inhibit signals, or the like, that occur withina certain time. The sense signal 350 could be analog or digital innature. In response to the sense signal 350, the feedback control block302 produces a feedback current I_(FB) 356 that corresponds to a peakswitching current required to maintain a regulated output. CurrentI_(FB) is converted to a voltage V_(FB) 352 by a resistor R1 358. Afixed current, proportional to a minimum peak switch current I_(PKMIN),is likewise converted to a fixed reference voltage V_(IPKMIN) 354 by aresistor R2 360. The voltage V_(FB) 352 and the fixed reference voltageV_(IPKMIN) 354 are received by a modulation block 306 to produce a pulsewidth modulated switching signal 308 with fixed switching frequencycorresponding to a fixed switching cycle period T_(S) at a first inputto an AND gate 310. In one example, block 306 changes the pulse width ofswitching signal 308 by adjusting the peak current I_(PK) in a switch S1120 of the power supply. Current in the switch S1 120 is converted to acurrent sense voltage V_(ISI) 372 by a current sensing resistor Rs 370.Current sense voltage V_(ISI) 372 is received by block 306. In oneexample, the peak current I_(PK) is proportional to the feedback currentI_(FB) 356 when the feedback current I_(FB) 356 is greater than aminimum reference peak switch current I_(PKMIN), and peak current I_(PK)is fixed at the value I_(PKMIN) when the feedback current is less thanI_(PKMIN). One example of a circuit that performs the function of block306 is shown in FIG. 3B and described later in this disclosure. Acomparator 314 compares voltage V_(FB) 352 to a reference voltageV_(IPKMIN) 354. In the example of FIG. 3A, the output of comparator 314is received as signal 316 by a multi-cycle modulator circuit 340 at theinput of an inverter 320 and to a first input of a NAND gate 324. Asshown in the example, multi-cycle modulator circuit 340 generates signal332, which enables or disables the switch signal 312 at the output ofAND gate 310 in response to the signal 316 from comparator 314. Signal316 from comparator 314 is high when voltage V_(FB) 352 is greater thanreference voltage V_(IPKMIN) 354. Signal 316 from comparator 314 is lowwhen voltage V_(FB) 352 is less than reference voltage V_(IPKMIN) 354.When signal 316 is high, multi-cycle modulator circuit 340 sets a signal332 high at a second input to AND gate 310, thereby enabling orpermitting the pulse width modulated switching signal 308 from block 306to appear uninterrupted at the output of AND gate 310, which in theillustrated example is also a switch signal 312 that is coupled to aswitch S1 120. In one example, switch S1 120 is a power metal oxidefield effect transistor (MOSFET). When signal 332 is low at the secondinput to AND gate 310, the pulse width modulated switching signal 308 isdisabled or not permitted to appear at the output of AND gate 310.

When signal 316 is low, period T_(S) is fixed at a value of T_(SG).Also, when signal 316 is low, the multi-cycle modulator circuit 340allows switching signal 308 to appear at switch signal 312 for a minimumnumber N_(MIN) of switching cycles, each having period T_(SG), anddisables switch signal 312 for a minimum number M_(MIN) of switchingcycles, each having period T_(SG). In one example, multi-cycle modulatorcircuit 340 includes a latch coupled to receive signal 316 from theoutput of comparator 314. As shown in the example, the latch is formedwith NAND gates 326 and 328, and inverters 320 and 330 and is coupled toan edge detector 348, which is coupled to a one-shot timer 338. Edgedetector 348 includes an XOR gate 342, inverter 344 and delay 346 asshown.

In one example, edge detector 348 produces logic low on line 336 forapproximately 30 nanoseconds when signal 332 from the output of inverter330 of the latch changes from high to low and from low to high. Whenline 336 goes low, one-shot timer 338 produces a logic low on line 334for a duration of either N_(MIN)T_(SG) or M_(MIN)T_(SG). The duration ofthe logic low on line 334 is N_(MIN)T_(SG) when line 332 is high. Theduration of the logic low on line 334 is M_(MIN)T_(SG) when line 332 islow. In one embodiment, N_(MIN) is chosen to be approximately 4 cyclesand M_(MIN) to be zero cycles. In one embodiment, the frequencycorresponding to fixed period T_(SG) is 30 kHz.

FIG. 3B shows one example of a circuit that performs the function ofblock 306 in of FIG. 3A, which includes a fixed frequency peak currentmodulator and a mode selector comparator. Mode select comparator 368receives voltages V_(FB) 352, V_(IPKMIN) 354, and V_(ISI) 372. Modeselect comparator 368 produces an output 374 that is received by oneinput of a latch 362. Another input of latch 362 receives a timingsignal 364 from an oscillator 366. The output of latch 362 is the pulsewidth modulated switch signal 308.

FIG. 4 is another example flow diagram that describes generally anothermethod to control a power supply according to the teachings of thepresent invention. Starting in block 405, sensing of power supplyoutput(s) occurs at block 410.

Next, in block 415, the controller uses information sensed from thepower supply output(s) to estimate an equivalent switching frequencyf_(EQ) required to regulate the output with period modulation. Next, theestimate of switching frequency f_(EQ) is compared to a referencefrequency f_(REF) at a decision block 420. In one example, the referencefrequency f_(REF) is greater than or equal to the highest audiofrequency of interest. In one embodiment, the reference frequency isapproximately 30 kHz.

The estimate of switching frequency f_(EQ) and comparison to referencefrequency f_(REF) do not need to be precise or the result of numericalcomputation. In one example, switching frequency f_(EQ) is estimated tobe higher or lower than reference frequency f_(REF) on the basis of afeedback signal. An estimate of f_(EQ) lower than f_(REF) implies alight load that demands a switching frequency within the range of audiofrequencies. Under those conditions, decision block 420 finds thatf_(REF)>f_(EQ), and the flow is directed to block 430. Period T_(S) isfixed at the value T_(SG) in block 430. Regulation is then accomplishedwith multi-cycle modulation in a block 435.

As shown in block 435, multi-cycle modulation accomplishes regulation ofthe output by gathering N cycles of switching in the switch signal, eachhaving period T_(SG), followed by no switching for M cycles of periodT_(SG). In the example, N and M are modulated, restricted respectivelyto minimum values N_(MIN) and M_(MIN). The resulting pattern ofswitching and no switching in the switch signal gives the same averagecurrent I_(D) 115 in FIG. 1A as uninterrupted switching at switchingfrequency f_(EQ) with a fixed switching cycle period. In the example,the pattern of switching with group modulation generates substantiallyless audio noise than an ungrouped pattern with ordinary periodmodulation in accordance with the teachings of the present invention.

If the decision block 420 finds that f_(REF)≦f_(EQ), then the flow isdirected to a block 425 so that regulation is accomplished with periodmodulation instead of multi-cycle modulation. In one embodiment, theperiod modulation frequency range is 30 kHz to 66 kHz. In anotherembodiment, period modulation frequency range is 30 kHz to 132 kHz. Inone example, multi-cycle modulation may be used to regulate the deliveryof low output power and period modulation may be used to regulate thedelivery of higher output power in accordance with the teachings of thepresent invention.

Thus, in one example, dependent upon the relationship between switchingfrequency f_(EQ) and reference frequency f_(REF), the switch may becontrolled to conduct according to a first or a second operating mode.In one operating mode, a switch is controlled to conduct within aswitching cycle and the period of the switching cycle is adjusted toregulate the output of a power supply. Alternatively, in a secondoperating mode, the switch may be controlled to conduct within aswitching cycle having a fixed period, one group of consecutiveswitching cycles separated from a next group of consecutive switchingcycles by a time of no switching, the time of no switching is adjustedto regulate the output of the power supply. In the example, the time ofno switching may be adjusted so that the frequency corresponding to theperiod of a switching cycle falls outside an audible frequency range. Inthe example, the group of switching cycles includes at least fourconsecutive switching cycles.

FIG. 5 illustrates generally an example schematic of a controller 545for a power supply that uses the method illustrated in FIG. 4. In theexample, a feedback control block 302 is coupled to receive a sensesignal 350 that is related to the output(s) of the power supply. Asdiscussed the sense signal 350 may represent a combination of one ormore parameters such as output voltage, output current, output power,current in the switch, timing of the switch, power supply input voltage,enable signals, inhibit signals, or the like, that occur within acertain time. The sense signal 350 could be analog or digital in nature.In response to the sense signal 350, the feedback control block 302produces an equivalent switching frequency signal f_(EQ) 504 thatcorresponds to a switching frequency required to maintain a regulatedoutput. The signal f_(EQ) 504 is received by a period modulation block506 to produce a period modulation switching signal 508 with period Tsat a first input to an AND gate 310. A comparator 514 compares signalf_(EQ) 504 to a reference signal f_(REF) 518, which in one examplecorresponds to a minimum switching frequency desired for periodmodulated switching. In the example of FIG. 5, the output of comparator514 is received as signal 516 by a multi-cycle modulator circuit 340 atthe input of an inverter 320 and to a first input of a NAND gate 324. Asshown in the example, multi-cycle modulator circuit 340 generates signal332, which enables or disables the switch signal 512 at the output ofAND gate 310 in response to the signal 516 from comparator 514. Signal516 from comparator 514 is high when signal f_(EQ) 504 is greater thanreference signal f_(REF) 518. Signal 516 from comparator 514 is low whensignal f_(EQ) 504 is less than reference signal f_(REF) 518. When signal516 is high, multi-cycle modulator circuit 340 sets a signal 332 high ata second input to AND gate 310, thereby enabling or permitting theperiod modulation switching signal 508 from period modulation block 506to appear uninterrupted at the output of AND gate 310, which in theillustrated example is also a switch signal 512. When signal 332 is lowat the second input to AND gate 310, the period modulation switchingsignal 508 is disabled or not permitted to be appear at the output ofAND gate 310.

When signal 516 is low, multi-cycle modulator circuit 340 allowsswitching signal 508 to appear at switch signal 512 for a minimum numberN_(MIN) of group switching cycles with period T_(SG), and disablesswitch signal 512 for a minimum number M_(MIN) of group switching cycleswith period T_(SG). In one example, multi-cycle modulator circuit 340includes a latch coupled to receive signal 516 from the output ofcomparator 514. As shown in the example, the latch is formed with NANDgates 326 and 328, and inverters 320 and 330 and is coupled to an edgedetector 348, which is coupled to a one-shot timer 338. Edge detector348 includes an XOR gate 342, inverter 344 and delay 346 as shown.

In one example, edge detector 348 produces logic low on line 336 forapproximately 30 nanoseconds when signal 332 from the output of inverter330 of the latch changes from high to low and from low to high. Whenline 336 goes low, one-shot timer 338 produces a logic low on line 334for a duration of either N_(MIN)T_(SG) or M_(MIN)T_(SG). The duration ofthe logic low on line 334 is N_(MIN)T_(SG) when line 332 is high. Theduration of the logic low on line 334 is M_(MIN)T_(SG) when line 332 islow.

In yet another example, a different parameter other than a peak switchcurrent I_(PK) or an equivalent switching frequency f_(EQ) could betaken as the criterion to choose between multi-cycle modulation and analternative mode of control. For example, the alternative mode ofcontrol may be a fixed switching frequency voltage mode PWM. In thismode on time of the switch is modulated as long as on time is abovecertain threshold in response to feedback signal. When the feedbacksignal demands an on time that is below a minimum threshold valuerepresenting a light load condition, the on time could be fixed at thisvalue and the mode of control can be switched to multi-cycle modulation.Yet another alternative mode of control may be a variable frequency witha fixed on-time or a fixed off-time that uses a minimum switchingfrequency as an indication for multi-cycle modulation.

Accordingly, in one example, a controller is coupled to a switch in apower supply to control the switch to switch on and off within aswitching cycle. In the example, the switch is coupled to an energytransfer element such as energy transfer element T1 125 of the powersupply. The group of switching cycles is separated from a next group ofswitching cycles by a time of no switching and the time of no switchingis adjusted to regulate a transfer of energy from an input of the powersupply to an output of the power supply. In the example, the minimumnumber of consecutive cycles in a group of switching cycles is fixed.Furthermore, the controller may operate under an additional operatingmode to regulate a higher output of power by controlling the switch toconduct within a switching cycle and adjusting the period of the cycleto regulate the output. To illustrate, FIG. 6, FIG. 7, FIG. 8, and FIG.9 show generally relationships between patterns of switching cycles andthe generation of frequencies that can produce audio noise to enableselection of values for the parameters for multi-cycle modulation inaccordance with teachings of the present invention.

Current I_(D) 115 produces a magnetic flux density B in the energytransfer element T1 125 (See FIG. 1). FIG. 6A is a graph showing oneexample of a typical waveform of magnetic flux density B as a functionof time. A peak flux density B_(P) occurs at the peak current I_(MAX) ofthe current I_(D) 115. The magnetic flux density B is active for a timeT_(A) within period T_(S).

It is important to consider the spectral content of magnetic fluxdensity B because the magnetic flux density produces mechanical forcesthat change the shape of energy transfer element T1 125. If magneticflux density B contains frequency components within the range of audiofrequencies that can cause the energy transfer element T1 125 toresonate, it can generate objectionably high audio noise.

FIG. 6B and FIG. 6C show partial frequency spectra of the waveform inFIG. 6A. Spectra from a high switching frequency are shown in FIG. 6B,whereas spectra from a low switching frequency are in shown in FIG. 6Con the same frequency scale.

The vertical lines on the frequency axes show the relative magnitudes ofcoefficients in the complex Fourier series representation of thewaveform in FIG. 6A. One skilled in the art will recall that a periodicfunction f(t) with period T_(O) may be represented as

${f(t)} = {{\sum\limits_{n = {- \infty}}^{\infty}{c_{n}{\mathbb{e}}^{j\; n\;\omega_{0}t}}} = {c_{0} + {\sum\limits_{n = 1}^{\infty}{2{c_{n}}{\cos\left( {{n\;\omega_{0}t} + \phi_{n}} \right)}}}}}$where$j = {{\sqrt{- 1}\mspace{14mu}{and}\mspace{14mu}\omega_{0}} = \frac{2\pi}{T_{0}}}$The coefficients c_(n) are in general complex numbers that have a realpart and an imaginary part. Although real and imaginary parts must beused for correct computations, only the magnitudes of the coefficientsc_(n) need be considered to explain and understand the invention.

In FIG. 6B and FIG. 6C it is not necessary to show the relativemagnitudes of the coefficients c_(n) for n<0 because the magnitudes atthe negative frequencies are the same as those at the positivefrequencies.|c _(n) |=|c _(−n)| for n=1,2, . . .The average of the magnetic flux density B in FIG. 6A is the coefficientc_(O), and is represented by the line at zero frequency in FIG. 6B andFIG. 6C. The coefficient of the fundamental frequency f_(S) thatcorresponds to the period T_(S) is represented by the line at frequencyf_(S) in FIG. 6B and FIG. 6C. The magnitudes of harmonics of thefundamental frequency are shown as lines at integer multiples of thefundamental frequency.

The force that produces mechanical distortion in a typical energytransfer element, such as a ferrite core transformer, to cause theproduction of audio noise is proportional to the square of the magneticflux density. It is useful to display the magnitudes of the coefficientsbecause the power at each frequency is also represented by the square ofthe magnitudes. Therefore, it is desirable to reduce the magnitude ofall coefficients within the range of audio frequencies.

To illustrate, FIG. 6B and FIG. 6C show a range of frequencies between alower boundary f_(XMIN) and an upper boundary f_(XMAX) where it isdesired to reduce or exclude all excitation from the magnetic fluxdensity in FIG. 6A. In one example, f_(XMIN) and f_(XMAX) arerespectively the lower and upper boundaries of the range of audiofrequencies that corresponds to the range of resonance frequencies ofcomponents such as transformers and ceramic capacitors that are commonlyused in switching power supplies. In one embodiment, the typical valueof f_(XMIN) is 8 kHz and that of f_(XMAX) is 15 kHz.

At the high switching frequency that produces the spectrum of FIG. 6B,there are no spectral components in the region between f_(XMIN) andf_(XMAX). As the period T_(S) is increased and the switching frequencygoes lower to regulate an output, the fundamental component atfundamental frequency f_(S) and all the harmonic multiples offundamental frequency f_(S) move closer to zero on the frequency axis.In the example of FIG. 6C, the fundamental component of the switchingfrequency has moved from a value greater than f_(XMAX) into theexclusion region between f_(XMIN) and f_(XMAX), which could cause, forexample, the transformer to resonate.

It is possible to rearrange the active times T_(A) of the waveform ofthe magnetic flux density in FIG. 6A to reduce the power contained inthe region between f_(XMIN) and f_(XMAX) while maintaining the sameaverage value required to regulate the output. One example of a suitablerearrangement is shown in FIG. 7A with its frequency spectrum in FIG.7B.

FIG. 7A shows a group of N switching cycle periods or N periods at aswitching frequency f_(SG) with corresponding period T_(SG), followed bya time of no switching. The time of no switching is a multiple M of theperiod T_(SG). The multiplication factor M is adjusted to keep the sameaverage magnetic flux density as the waveform of FIG. 6A that would beobtained with an equivalent switching frequency f_(EQ). The group periodT_(G) is the sum of the N periods T_(SG) and the time of no switching.Thus, to regulate an output, the period T_(SG) is unchanged while thegroup period T_(G) is adjusted.

In this description we refer to the adjustment of the period T_(S) of aswitching cycle in FIG. 6A as period modulation. We refer to theadjustment of the group period T_(G) in FIG. 7A as multi-cyclemodulation.

FIG. 7B shows an example spectrum drawn to scale of the group modulatedmagnetic flux density of FIG. 7A for the conditions of N=M=11 andT_(SG)=2T_(A). FIG. 6C shows the spectrum with period modulated magneticflux density that produces the same output power as the spectrum in FIG.7B with group modulated magnetic flux density. The switching frequencyor frequency f_(S) in FIG. 6C is the equivalent frequency f_(EQ) in FIG.7B. Comparison of the spectra in FIG. 6C and FIG. 7B shows thatmulti-cycle modulation greatly reduces the power within the exclusionrange between f_(XMIN) and f_(XMAX). The total power within a range offrequencies is represented by the sum of the squares of the magnitudeswithin the range.

To further illustrate, the multi-cycle modulated magnetic flux densityof FIG. 7A can be viewed as the unmodulated signal in FIG. 8A multipliedby the multi-cycle modulator signal of FIG. 8B. Design parameters formulti-cycle modulation can be determined by examination of the spectraof the two signals.

FIG. 8C shows the relative magnitudes of the coefficients for positiveand negative frequencies in the complex Fourier series representation ofthe multi-cycle modulator signal of FIG. 8B. The case of N=M is takenfor ease of illustration since the even numbered harmonics are zero. Oneskilled in the art can easily determine the coefficients for any valuesof N and M.

FIG. 9 shows how the spectrum of the unmodulated switching signal ofFIG. 8A and the spectrum of the multi-cycle modulator signal of FIG. 8Bare combined to produce the spectrum of the multi-cycle modulated signalof FIG. 7A. One skilled in the art will recall that the multiplicationof two signals in the time domain is equivalent to the convolution ofthe spectra of the two signals in the frequency domain. That is, themultiplication of the unmodulated magnetic flux density signal B(t) andthe group modulator signal G(t) produces the group modulated signal C(t)that has the spectrum c(f) such thatc(f)=∫_(−∞) ^(∞) b(x)g(f−x)dxwhere b(f), g(f), and c(f) are the spectra of B(t), G(t), and C(t)respectively, and x is a dummy variable of integration along thefrequency axis.

The convolution operation is easily interpreted and performedgraphically, especially when the spectra are nonzero at only discretefrequencies. To perform the graphical convolution, first flip or reversethe spectrum of the group modulator about the zero frequency axis sothat frequencies get more positive to the left and more negative to theright. Although the reversed spectrum of magnitudes is identical to theoriginal spectrum because of symmetry, the reversal is necessary forcorrect numerical computations that require the proper signs in thecoefficients.

In FIG. 9, the positive frequency portion of the spectrum of theunmodulated switching waveform is shown for the case of T_(SG)=2T_(A).Only the coefficient at zero frequency and the fundamental frequency areshown because the higher harmonics are much greater than the highestfrequency of interest. A graphical procedure may be used to obtain themagnitude of the spectrum of the multi-cycle modulated switchingwaveform at any frequency of interest. To obtain the coefficients forthe spectrum of the multi-cycle modulated switching waveformgraphically, slide the reversed spectrum of the multi-cycle modulatoralong the frequency axis such that the zero of the multi-cycle modulatorspectrum is at the frequency of interest. Then multiply each coefficientin the multi-cycle modulator spectrum by the coefficient of theunmodulated spectrum that aligns with it. Sum all the products to obtainthe value of the coefficient for the multi-cycle switching spectrum atthe frequency of interest.

Although the spectrum of the multi-cycle modulator contains an infinitenumber of coefficients, only the most significant need to be consideredto obtain a good approximation of the multi-cycle switching spectrum. Inmost practical cases, all coefficients beyond the first three or fourcan be ignored. The example in FIG. 9 shows that the magnitude of thecoefficient c₁₅ in the multi-cycle modulated switching spectrum isapproximately the product of the magnitude of the coefficient b₁ of theunmodulated switching signal and the magnitude of the coefficient g₇ ofthe multi-cycle modulator. The value is approximate because the productsof all higher harmonics are ignored. In the example of FIG. 9, thecoefficients at the even numbered frequencies are exact because the sumcontains only one term that is not zero.

Study of FIG. 9 yields relationships to guide the selection ofparameters for group modulation. With the region of excluded frequenciesdetermined by f_(XMIN) and f_(XMAX), where f_(XMIN)<f_(XMAX), the groupswitching frequency f_(SG) is chosen such thatf _(SG) ≧f _(XMIN) +f _(XMAX)determines the highest significant frequency in the spectrum of themulti-cycle modulator as a multiplier W of the fundamental frequencyf_(G) such that all frequencies greater than W multiplied by f_(G)should be negligible in the convolution that determines the coefficientsin the region of excluded frequencies. Typically, only the first two orthree harmonics of f_(G) will have any real significance, so themultiplier W is usually not greater than 2 or 3. Then

$N \geq {W\frac{f_{XMAX}}{f_{XMIN}}}$optimum values are the lowest required for sufficient reduction in theregion of excluded frequencies.

For an equivalent frequency f_(EQ) that produces the same output powerwith multi-cycle modulation as with period modulation, the modulationmultiplier M must be

$M = {N\left( {\frac{f_{SG}}{f_{EQ}} - 1} \right)}$corresponding to a multi-cycle group frequency

$f_{G} = \frac{f_{EQ}}{N}$In examples where the time between groups must be an integer multiple ofT_(SG), the correct value of M is satisfied as an average over severalgroup periods with different integer values of M. In one design, it maybe desired to set a minimum integer value of M such that M≦N. Since M isminimum when f_(EQ)=f_(XMAX), choose f_(SG)=2f_(XMAX). With theconstraint of minimum M, a choice of W=3 is usually valid for all M. Asan example, a typical design to reduce frequencies in the audio rangebetween f_(XMIN) of 8 kHz and f_(XMAX) of 15 kHz where a transformer islikely have mechanical resonances may use f_(SG)=30 kHz. A choice of W=3would give N≧5.625 so N=M=6. Regulation with group modulation wouldgroup together a minimum of six periods of consecutive switchingfollowed by a minimum of six periods of no switching.

Thus, in one example, a switch in a switching power supply is controlledto switch on and off within a switching cycle of substantially fixedperiod. At least one group of switching cycles is separated from a nextgroup of switching cycles by a time of no switching and the time of noswitching is adjusted to regulate an output of the switching powersupply. In the example, the time of no switching may be adjusted to atime that is an integer multiple of the substantially fixed period ofthe switching cycle. Additionally, the time of no switching may be atleast as long as the group of switching cycles.

The above description of illustrated examples of the present invention,including what is described in the Abstract, are not intended to beexhaustive or to be limitation to the precise forms disclosed. Whilespecific embodiments of, and examples for, the invention are describedherein for illustrative purposes, various equivalent modifications arepossible, as those skilled in the relevant art will recognize. Indeed,it is appreciated that the specific voltages, currents, frequencies,power range values, times, etc., are provided for explanation purposesand that other values may also be employed in other embodiments andexamples in accordance with the teachings of the present invention.

These modifications can be made to examples of the invention in light ofthe above detailed description. The terms used in the following claimsshould not be construed to limit the invention to the specificembodiments disclosed in the specification and the claims. Rather, thescope is to be determined entirely by the following claims, which are tobe construed in accordance with established doctrines of claiminterpretation.

What is claimed is:
 1. A controller for use in a power supply, thecontroller comprising: a switch signal generator coupled to generate aswitch signal to control switching of a switch of the power supply; amodulation circuit coupled to provide the switch signal generator with amodulation switching signal responsive to a switching frequency signalthat represents an equivalent switching frequency of the switch at whichan output of the power supply would be in regulation, wherein themodulation circuit modulates the period of the modulation switchingsignal to change the switching period of the switch signal in responseto the switching frequency signal indicating that the equivalentswitching frequency is greater than a reference frequency, and whereinthe modulation circuit fixes the period of the modulation switchingsignal to fix the switching period of the switching signal in responseto the switching frequency signal indicating that the equivalentswitching frequency is less than the reference frequency; a multi-cyclemodulator circuit coupled to the switch signal generator, wherein themulti-cycle modulator circuit enables the switch signal generator toprovide the switch signal uninterrupted in response to the switchingfrequency signal indicating that the equivalent switching frequency isgreater than the reference frequency, and wherein the multi-cyclemodulator circuit disables the switch signal generator for a first timeperiod and then enables the switch signal generator for a second timeperiod in response to the switching frequency signal indicating that theequivalent switching frequency is less than the reference frequency,wherein the multi-cycle modulator varies the first time period toregulate the output of the power supply.
 2. The controller of claim 1,wherein the multi-cycle modulator circuit includes a timer coupled tomaintain the first time period for a first minimum time periodindependent of the switching frequency signal.
 3. The controller ofclaim 1, wherein the multi-cycle modulator circuit includes a timercoupled to maintain the second time period for a second minimum timeperiod independent of the switching frequency signal.
 4. The controllerof claim 1, wherein the switch signal generator is an AND gate having afirst input coupled to receive the modulation switching signal, a secondinput coupled to receive a signal from the multi-cycle modulator circuitto enable and disable the AND gate, and an output to provide the switchsignal.
 5. The controller of claim 1, further comprising: a feedbackcircuit to be coupled to receive a sense signal related to the output ofthe power supply regulator and coupled to generate the switchingfrequency signal in response thereto; and a comparator coupled to thefeedback circuit to compare the switching frequency signal with a signalrepresentative of the reference frequency, wherein the multi-cyclemodulator circuit is coupled to an output of the comparator.
 6. Thecontroller of claim 5, wherein the sense signal represents a combinationof one or more parameters selected from the group consisting of: outputvoltage of the power supply, output current of the power supply, outputpower of the power supply, current in the switch, timing of the switch,power supply input voltage, enable signals, inhibit signals.
 7. Thecontroller of claim 5, wherein the equivalent switching frequency beinglower than the reference frequency implies a light load at the output ofthe power supply that demands a switching frequency within a range ofaudio frequencies.
 8. The controller of claim 1, wherein the first timeperiod is an integer multiple of the fixed switching period.
 9. Thecontroller of claim 8, wherein the second time period is an integermultiple of the fixed switching period.
 10. The controller of claim 1,wherein the multi-cycle modulator adjusts the first time period suchthat a frequency of a plurality of switching cycles falls outside anaudible frequency range.
 11. The controller of claim 10, wherein thereference frequency is greater than or equal to a highest audiofrequency of interest.
 12. The controller of claim 11, wherein thereference frequency is approximately 30 kHz.
 13. The controller of claim11, wherein the modulation circuit modulates the period of themodulation switching signal to change the switching period of the switchsignal such that a lower end of a frequency range of the switchingsignal is the highest audio frequency of interest.
 14. The controller ofclaim 13, wherein the frequency range of the switching signal, whenmodulated by the modulation circuit, is about 30 kHz to about 66 kHz.15. The controller of claim 13, wherein the frequency range of theswitching signal, when modulated by the modulation circuit, is about 30kHz to about 132 kHz.